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Posted 5/28/98

James Miller, G3RUH, has been a prolific writer and very hard worker in the radio amateur digital communications field. This page presents three of his now classic papers on "modern" digital communications. James presents his ideas in such a down to earth, practical manner that I think it is appropriate to reprint these papers here as tutorial information.   Modern digital communication system designers can learn a lot from James' ideas and approach to practical communications theory.

All of this work is the sole copyright property of James Miller, G3RUH, and this information can ONLY be used for educational or ham radio related, non-commericial applications.  We thank James a great deal for allowing us to reprint this valuable tutorial here in SSS Online.

Shannon, Coding and the Radio Amateur (Part I)

The Shape of Bits To Come (Part II)

9600 Baud Packet Radio Modem Design (Part III)

A Concrete Modem Design Example:

ABSTRACT: 9600 Baud Packet Radio Modem Design
Papers of ARRL 7th Computer Networking Conference (US), Oct 1988. pps 135- 140

Also: Proceedings of the first RSGB Data Symposium, Harrow, England July 1988. (12 pps).

Also: "Packet: Speed, More Speed and Applications", ARRL 1995. ISBN 0-87259-495-5 Chapter 1.

The theoretical minimum audio bandwidth required to send 9600 baud binary data is 4800 Hz. Since a typical NBFM radio has an unfiltered response from zero to some 8 kHz, transmission of 9600 baud binary data is perfectly possible through it. This paper describes a successful implementation.

It is available on-line from as the file: /amsat/articles/g3ruh/

This placement: /amsat/articles/g3ruh/
Date: 1995 Mar 29

First placement: Papers of ARRL 7th Computer Networking Conference (US), Oct 1988. pps 135-140

Other placements:
  • Proceedings of the first RSGB Data Symposium, Harrow, England July 1988. (12 pps).

  • Packet: Speed, More Speed and Applications, ARRL 1995. ISBN 0-87259-495-5
Corrections: Minor edits and update 1994
Size: 22,000 bytes 3400 words 450 lines


9600 Baud Packet Radio Modem Design

James Miller BSc, G3RUH
3 Benny's Way

The theoretical minimum audio bandwidth required to send 9600 baud binary data is 4800 Hz. Since a typical NBFM radio has an unfiltered response from zero to some 8 kHz, transmission of 9600 baud binary data is perfectly possible through it. This paper describes a successful implementation.


The standard packet VHF/UHF radio data rate is 1200 baud because all TNCs provide an internal modem for this speed, and the two-tone AFSK audio spectrum suits unmodified voiceband radios comfortably. However all TNCs can generate much higher data rates, and most FM radios have an unrealised audio bandwidth of some 7-8 kHz or more. So in many cases 9600 baud radio transmission is entirely practical with them.

This design is a high performance full duplex modem designed for packet use with most voiceband NBFM radios, assuming only minor modifications.

A key feature of this modem is its digital generation of the transmit audio waveform. Precise shaping compensates exactly for the amplitude and phase response of the receiver. This results in a "matched filter" system, which means that the received audio offered to the data detector has the optimum characteristic ("eye") for minimum errors. It also allows very tight control of the transmit audio bandwidth.


Here is a summary of the modem features.
  • MODULATION: FM. Audio applied direct to TX varactor. +/- 3 kHz deviation gives RF spectrum 20 kHz wide (-60db). Fits standard channel easily.

  • TX MODULATOR: 8 bit long digital F.I.R. transversal filter in Eprom for transmit waveform generation (12 bit optional). Gives "brick wall" audio spectrum. Typically -6 db at 4800 Hz, -60db at 7500 Hz. Allows compensation for receiver (the channel) to achieve perfect RX "eye". Up to 32 TX waveforms, jumper selectable. Output adjustable 0-8v pk-pk.

  • SCRAMBLER (Randomiser): 17 bit maximal length LFSR scrambler, as per K9NG system, and UoSAT-14/22/23 etc. Jumper selectable Data or BERT (bit error rate test) mode.

  • RX DEMODULATOR: Audio from receiver discriminator, 10mv-10v pk-pk. 3rd order Butterworth filter, 6 kHz. Data Detect circuit for use on simplex (CSMA) links. Independent un-scrambler.

  • CLOCK RECOVERY: New digital PLL clock recovery circuit with 1/256th bit resolution. Average lock-in time 50 bits (depends on SNR).

  • CONNECTS to AX.25 TNC "Modem Disconnect" jack. Suitable for TNC-2 and any other provided the internal modem can be bypassed. Standard TNC digital connections needed: TXData, TXClock (16x bit rate), RXData, Data Detect DCD, GND. RXClock available. RADIO: TXAudio, RXAudio, GND, All connections via 0.1" pitch pads for SIL connectors or direct soldering. Unwired DIN 41612 96-way connector (use optional).

  • POWER CONSUMPTION: 10 - 15 v DC at 40ma (CMOS Roms), 170 ma (NMOS Roms). Total 19 ICs (13 CMOS, 2 DACs, 2 op-amps, 2 Eproms). 5 volt regulator and heatsink.

  • OTHER FEATURES: The only set-up is TXAudio level. Channel calibration facility. Audio loopback. No hard-to-get parts.

  • PCB: 160x100mm (single Eurocard format). Top professional quality, double sided, maximum copper ground plane, plated through, solder resist, yellow silk-screen. Four 3.3mm mounting holes.


This modem is obviously only suitable for a TNC if its internal modem can be bypassed, and if it provides for the TTL digital signals:

 * TXData                       e.g. TNC-2 J4-19
 * TXClock (16x data rate)                 J4-11
 * RXData                                  J4-17
 * Data Detect ("DCD")                     J4- 1
 * GND                                     J4-15
 * RXClock (optional)                     not used
TAPR TNC-2 based designs do this, typified by the TNC-2, PK-80, MFJ-1270, TNC-200. Close relatives (but with minor variations) are Tiny-2, Euro TNC2C, BSX-2, PK-87, PK-88 and TNC-220. The necessary interface is at the "modem disconnect" jack.

The modem's use is not confined to TNCs, however. Some of the recent multiport packet switches, indeed any signalling system, is suitable if it can service the minimum signal set above.


The ideal would be to have a flat DC-8 kHz radio link. The better the TX and RX specification, the better the received data at the detector, and hence less susceptibility to errors.

Some apparently horrid receiver responses still offer useable service, but with a typically 3 db reduction in performance. A good radio achieves about 1.5 db implementation loss compared with a perfect link.

Remember that one is pushing most radios to their limit since they were designed for speech where even 100% distortion is still intelligible. A little more finesse is required for data transmission.

  • NBFM design
  • Output from discriminator (essential)
  • Response to DC (virtually essential)
  • Response no worse than -4 db at 4.8 kHz
  • No worse than -10 db at 7.2 kHz
  • As smooth/flat a phase delay as possible
  • As smooth an amplitude response as possible
  • Little change in response with 2 kHz de-tuning off-channel.
On the whole, most receivers will perform as required. Those with the least complicated IF filtering appear best, especially those with type "D" 20 kHz channel filters (e.g. CFW455D), though the "E" (16 kHz) is OK too.

Radios with dozens of tuned circuits tend to be fussy, and should be carefully aligned for even response, decent linearity, phase delay and mistuning performance.

  • MUST generate true FM
  • Response DC to 7.2 kHz (essential).
Transmitters based on Xtal oscillator/multipliers are likely to be the most appropriate. (Usually base stations. So who wants to tie up a multimode radio on a link anyway!)

Transceivers (synthesised or not) that have quite separate oscillator sub- systems for generating FM and possibly SSB/CW, which is then mixed with a synthesised source to produce the final carrier are OK.

Simpler synthesised FM transmitters, where the varactor modulated oscillator is within the synthesis PLL are generally not useable, as the PLL tracks the modulation, and so you get no LF response, There are ways around this by modulating the reference xtal, called two-point modulation.

Remember you need true FM, which preferably means a varactor pulling the oscillator frequency, NOT phase modulating a tuned circuit.


All the bits and pieces required to interface digital data to a radio are called a "modem", short for modulator/demodulator. These two functions are complementary, and essentially separate even though there may be shared parts such as clocks and power supplies etc. Figure 1a-1d is the full circuit diagram of production boards, issue 3. Issue 1 were prototypes, issue 2 the beta-test models.

Fig 1a

Fig 1a

Fig 1a

Fig 1a
Fig. 1a-d. Modem schematic ; split into 4 parts for legibility.


Data for transmission is first passed through a randomiser or scrambler. This ensures that there are no long runs of all "1"s or "0"s or repeated patterns. There are several good reasons for doing this [2].

One is that the channel is not DC coupled. It could never be so in an FM system unless one could guarantee both transmitter and receiver were always exactly on frequency and had no drift. As this is virtually impossible to achieve, one simply AC couples the channel, i.e. gives it a response down to a few Hz, and exploits the feature of the randomised data that it has a negligible DC component.

Secondly, since the data stream is now randomised, its spectral energy is evenly spread out at all times. Intense spectral lines do not suddenly appear and create sporadic splatter into nearby channels.

A third reason is that since the data is guaranteed to have a regular supply of ones and zeros, the receiver's bit clock recovery and demodulation circuits work better.

Not surprisingly a burst of data sounds like a burst of radio noise, and is quite hard to distinguish from the unsquelched background.


The transmit waveshape(s) are stored in an EPROM. An 8 bit shift register contains the most recent bits, which are used to look up profile for the middle one. Four samples/bit go to make up the profile. In this way the transmitted waveform is synthesised not only from the present bit's state, but also four that preceded it and four to come.

The 8 bit value output from the EPROM is converted to a voltage by an inexpensive single-rail DAC, and is then analogue low pass filtered to remove harmonics of the clock and associated discrete phenomena. This is variously called "anti-aliasing" or "smoothing" or "interpolating". Either way, it simply joins up the dots!

The arrangement as a whole is a "finite impulse response filter", or FIR for short.

The Transmit EPROM is normally a type 27C128 and can hold between sixteen 8- bit long FIRs, to one 12-bit long FIR or various combinations. A 27C256 can also be used offering up to 32 responses. NMOS roms are also suitable.


Audio from the receiver discriminator is passed through a gentle input filter which removes out of band spurious noise, particularly IF residue. The signal is then limited and detected by sampling at the correct instant.


The detected data, still randomised is then passed through an unscrambler, where the original data is recovered, and this goes off to the TNC. A scrambler is very simple, consisting of a 17 bit shift register and 3 Exor gates. See for example fig 3 of [2].

The scrambling "polynomial" is 1 + X^12 + X^17. This means the currently transmitted bit is the EXOR of the current data bit, plus the bits that were transmitted 12 and 17 bits earlier. Likewise the unscrambling operation simply EXORs the bit received now with those sent 12 and 17 bits earlier. The unscrambler perforce requires 17 bits to synchronise.

This polynomial was deliberately chosen to be the same as implemented by Goode [1] in an earlier modem design. It will also be used on one of the UoSAT-C satellite downlinks.


A particularly useful by-product of scramblers is "bit error rate testing" or BERT for short. Suppose the transmitted data is held to all "1"s. Then a receiver's error-free output should also be all "1"s even though the transmitted data is quite random. So to test the quality of a link one merely sends all "1"s and attaches a counter at the other end.

If one bit is corrupted due to channel noise, the error will in fact appear exactly 3 times at the receiver output, because there are 3 versions of the scrambled stream exored together. Even though one error creates two more, this doesn't matter because just the single error is enough for a packet to be rejected.

Incidentally, randomisers/scramblers don't really violate rules concerning codes and ciphers any more than do ASCII, Baudot or Morse. Since the scrambling algorithm is freely published, the meaning of the data is not obscured.


The demodulator must extract a clock from the received audio stream. It's needed to time the receiver functions, including the all-important data detector.

The familiar TAPR TNC-2 state machine is not satisfactory in this application, as its resolution is only 1/16th bit. It can show jitter up to +/- 5/16ths of a bit in this narrow band application, which gives bad performance for detector timing.

This modem uses a new digital phase locked loop (DPLL) with a resolution of 1/256th bit.

The received audio is limited, and a zero crossing detector circuit generates one cycle of 9600 Hz for each zero-crossing (a proto-clock). This is compared with a locally generated clock in a phase detector based on an up/down counter. The counter increments if one clock is early, decrements otherwise. This count then addresses an Eprom in which 256 potential clock waveforms are stored, each differing in phase by 360/256 degrees. In this way the local clock slips rapidly into phase with that of the incoming data.

RX Clock lock-in time depends on the signal to noise ratio, and the initial phase error. A signal that's already in phase pulls into lock within 0 bits. A noise-free signal exactly out of phase will pull in to a point where data errors cease in about 80 bits. A very noisy signal could take up to 200 bits. In practice, an average figure is around 50 bits, or about 5 ms at 9600 baud.

Proto-clock and local clock are also compared in an exor gate, and when they are "in-phase", a Data Carrier Detected signal (DCD) is sent to the TNC. High or low options are available.


As mentioned, a strength of this modem is its digital generation of the transmit audio waveform. The precise shaping compensates exactly for the amplitude and phase response of the receiver. It also allows very tight control of the transmit audio bandwidth.

The waveform is synthesised as follows. First the "ideal" receiver output waveform (at the "eye point") is defined. This waveform, for one isolated bit, has a perfect "eye". It has a value of +1 at T=0 and a value of 0 at +/-T, +/-2T etc where T is a bit period. Its spectrum is flat to 3300 Hz, - 6db down at 4800 Hz, and is absolutely band limited to 6300 Hz. The waveform is called a "Nyquist Pulse".

Next the channel frequency and phase response is measured. It is made up of several contributions; the modem transmit anti-alias filter, the radio transmitter response, the receiver response and finally the modem receive filter. In practice most of these components are already characterised, so it's only necessary to characterise the receiver part explicitly.

Now the ideal Nyquist pulse's frequency response is divided by the channel frequency response to give the ideal transmit spectrum. This is then Fourier transformed to the time domain, and specifies EXACTLY the waveform of a transmitted bit that would pass right through the system to emerge with the desired Nyquist shape.

This desired waveform will have a time span of some 15 bits or more duration. However only the middle 8-12 bits duration will have any significant amplitude. So the extremes are gracefully smoothed off to exactly 8 bits span, a process known as windowing.

As a verification check, the new pulse is now "sent" mathematically forwards through the channel to assay the effect of having had to truncate it, and the "eye" point vertical jitter calculated. It is typically +/- 10% of a unit bit amplitude.

This calculation only defines a waveshape for one isolated bit. But it will extend over 8 bits elapsed time. So the final stage in the synthesis is to add up the impulse responses of all possible 256 combinations of preceding and trailing bits to give the true "convolved" waveform that is finally used. The waveform is then stored as numbers in an Eprom.

The software to do the equalisation is programmed in BASIC, except for the Fourier transforms (512 point complex FFTs) which are machine coded for speed. The waveforms and spectra are displayed graphically at every stage. Figures 2 a-f illustrate equalisation of a fairly extreme specimen of radio, a Midland 13-509 220 MHz transceiver (USA).

Fig 2a

Fig 2a. Frequency response and phase delay of a typical NBFM receiver. Note that the frequency axis is normalised. f=1.0 corresponds to 9600 Hz.

Fig 2b

Fig 2b. Impulse response corresponding to fig 2a. Horizontal axis time "ticks" are at intervals of 1 bit (1/9600 sec), and has been shifted some 125 us so as to centre the peak. An isolated, unequalised bit would emerge from the receiver shaped like this. A stream of bits would be the sum of many of these. Bad features such as significant non-zero values at T = -2, +1 and +2 bits would give rise to substantial inter-bit interference and a very poor eye, making error free communication impossible. The purpose of equalisation is to eliminate this phenomenon.

Fig 2c

Fig 2c. Equalised Transmit bit. If the transmitter sends an isolated bit shaped like this, the receiver will give an output like fig 2e. This is the transmit waveform for one isolated bit. If this corresponds to a binary "1", a binary "0" is a negative pulse like this.

Fig 2d

Fig 2d. Spectrum of the transmitted audio pulse of fig 2c.

Fig 2e

Fig 2e. Receiver output response to an isolated bit which has been properly equalised. Note that generally zero crossings at the bit ticks (excepting T=0) are zero. This is the "Nyquist Pulse". In fact the equalisation is not perfect (e.g. at T=4). This is due to the fairly extreme radio response chosen for illustrative purposes.

Fig 2f

Fig 2f. The convolution of many bits of fig 2e superimposed looks like this, an "EYE" diagram showing a few hundred bits over 3 bit periods. You would see this on an oscilloscope. The data detector samples this waveform at the widest part of the eye. See how as a result of equalisation the eye is wide open, giving maximum noise immunity. Vertical spread at the sample point is mainly due to the aberration at T = 4 in fig 2e. Note also the considerable spread in zero crossing instants typical of narrow bandwidth systems, which lead to the need for careful clock recovery circuit design.


Users accustomed to the speed of a typical 1200 baud channel are usually astonished to see 9600 baud data scrolling off the screen at full speed.

In the perfect environment of audio loopback, the error performance of the modem has been measured, and the implementation loss is about 1 db. In other words the modem itself does not introduce significant degradation in system performance.

On the other hand it is not very appropriate to describe the modem in terms of microvolts at an FM receiver input. So much depends on the particular specimen, the environmental noise level, frequency drift, and goodness of equalisation. Steve Goode [1] has given detailed results for one typical situation, and it should be clear from this that there can be no simple snap performance measure, other than "does it work".

What can be said is that once the received signal strength puts the receiver output above the "spitching" threshold, and into smooth noise, (and that's only a 1db spread in RF level!) the 9600 baud system becomes essentially error free. So a radio link needs to be just over the noise threshold for good performance - as also does 1200 baud AFSK.

Remember, the purpose of this modem is to provide a reliable high speed communications facility, not to scrape weak DX off the noise floor!


The standard transmit Eprom contains waveforms for 16 named receivers. One of these (usually selection no. 10) is almost certain to be satisfactory for an arbitrary receiver. If not, the author offers a customising service. Full details are included in the modem Instruction Booklet. New receivers will gradually be added to the standard Eprom as users supply more data.


This project is supported with a Printed Circuit Board and full instructions. At the time of writing (88 Aug 16) some 200 are in worldwide use. (Update 1994 Nov: about 15,000).


PCB £18 post paid UK/Europe, £19 air-mail elsewhere. CMOS TX and RX Eproms (programmed), when ordered at the same time as the PCB, £6/pair. DACs £5/pair. Built and tested PCB £65. Any eproms ordered separately £5 each chip. You are free to copy the eproms if you wish.

Sterling Cheques, Eurocheques (max £150), Travellers Cheques, Cash, or bank draft drawn on a London bank. Also electronic funds transfer, please add £6 bank charges. No credit cards. You can "buy" English pound notes at many banks. I will also accept US dollars in cash only (green notes/ travellers cheques) at a rate of $2 per pound. In case of difficulty, contact me. Prices include postage and packing.

Note: £ = GBP = Pounds Sterling.

James Miller G3RUH, 3 Benny's Way, Coton, Cambridge, CB3 7PS, England
Tel: +44 1954 210388 Fax: +44 1954 211256


The modem design is also incorporated in products from:

PacComm Inc: NB-96
Kantronics: DE-9600
MFJ: MFJ-9600
Tasco: TMB-965
Symek: TNC2-H

and about a dozen more licensees.


Scores of people provided feedback and support for this project. These include some 20-25 beta testers world wide who provided many responses for the eproms, design criticism and debugged all those wretched non-standard "standard" modem disconnect headers! Gwyn Reedy W1BEL of PacCom Inc and Phil Bridges G6DLJ of Siskin electronics Ltd have been instrumental in promoting the design at both amateur and commercial levels. I also acknowledge the pioneering work of Steve Goode [1] who did a lot of spade work some years ago, and which convinced me all along that this project was on sound ground. Thanks too to Bob McGwier N4HY for insisting this design should see the light of day, and not remain the secret weapon of EastNet- UK's links.

  1. GOODE S. "Modifying The Hamtronics FM-5 for 9600 bps Packet Operation", Proceedings of the Fourth ARRL Amateur Radio Computer Networking Conference, pps 45-51.

  2. HEATHERINGTON D.A. "A 56 Kilobaud RF Modem", Proceedings of the Sixth ARRL Amateur Radio Computer Networking Conference, pps 65-75.

©1988 James Miller G3RUH         E-mail:

Higher Speeds with the G3RUH 9600 baud Packet Radio Modem
by James Miller G3RUH

1993 Aug 23

The modem is capable of speeds up to 64000 baud. This limit is set by the maximum rate that the DAC chips can operate. This note describes how to achieve rates from 4800 to 64000 baud. The slowest speed is suitable for 12.5 kHz channelised radios. The highest speed suits radios that have broadcast FM bandwidth filters.

To implement a higher speed you need to:
  1. Increase your TXData rate (!)
  2. Increase the associated TXClock
  3. Change some analogue filter components proportional to the speed increase.
It is not necessary to change either of the eproms. If you are going for a higher speed, it is likely that the radios involved are "specials" and you will already have wide bandwidth and flattish group delay, so the loopback selection 0 from the standard ROM will be OK.

The table below suggests the best conditions for different speeds. Component references are for my own PCB card. Clones are different.

                          Data Rate - Baud
 Comp         4800    9600   19200    38400   64000
 R6           220k    100k     47k     22k    15k

 R16          100k    100k    100k     47k    15k
 R17           82k     82k     82k     39k    12k
 R18           39k     39k     39k     18k     5k6
 R19           27k     27k     27k     15k     3k9
 R21          100k    100k    100k     47k    15k
 R22           56k     56k     56k     27k     8k2

 C18           4n7     4n7     4n7     1n    680p
 C20         220p    100p      47p    22p     12p

 C26     (   470p    220p     100p   100p    220p    )
 C27     (     2n2      1n    470p   470p      1n    )
 C28     (     2n2      1n    470p   470p      1n    )
 C29     (     6n8      3n3    1n5   470p    470p    ) 2% or
 C30     (   220p     100p     47p    47p    100p    ) better
 C31     (     1n     470p    220p   220p    470p    )
 C32     (     2n2      1n    470p   220p    150p    )
 Deviation +/- 1.5      3       6      12     20 kHz ) In FM
 IF Bandwidth    8     15      30      60    100 kHz ) Service
These modifications have been tested in both amateur and commercial service. All comments gratefully received, and added to the database.

73 de James G3RUH @ GB7DDX.#22.GBR.EU 1993 Aug 23 [Mon] 0917 utc

G3RUH 9600 Baud Packet Radio Modem PCB

SECTION 7 - COMPONENTS LIST From G3RUH PCB Manual (18 pages)

(See Section 12. for 19200 baud changes)

RESISTORS - (All 2%, 0.4" pitch)
R1,6,14-16,21      100k  (6)    R8      47k     R22     56k
R2,5                 4k7 (2)    R17     82k     R23      3k3
R3                  18k         R18     39k     R24     12k
R4,9-13,20,27-28    10k  (9)    R19     27k     R25,26  100  (2)
R7                  33k          Note:  4k7 = 4700 ohms, etc

VR1     10k  Cermet Trimmer e.g. Spectrol 63, 64 series,
             Bourns 3386, 3266, 3296 etc.  PCB Drilled for all types.

 C1-16,19,21-22,24-25   100n    20% monolithic (21)    0.1" pitch (NOT 0.2")
 C17,33-34      10u     16v tantalum   (3)             0.2" pitch
 C18        4n7     20% monolithic ceramic             0.1" pitch
 C20        100p    20% monolithic ceramic             0.1" pitch
 C23        1n      20% monolithic ceramic             0.1" pitch
 C26        220p    2.5% polystyrene                Note:
 C27-28,32  1n      2.5% polystyrene  (3)          100n  = 0.1u
 C29        3n3     2.5% polystyrene                10n  = 0.01u
 C30        100p    2.5% polystyrene                 4n7 = 0.0047u
 C31        470p    2.5% polystyrene                 1n  = 0.001u etc

 U1,3   4029    (2)     Up/down counter
 U2     27C64           Eprom - RX (or 27C128 and/or NMOS)
 U4,7,12,14,18  74HC164 (5)     8-bit shift register
 U5,17  74HC74  (2)     Dual D type
 U6,13  74HC86  (2)     Quad EXOR
 U8     74HC161         4-bit counter
 U9,19  ZN429E-8 (2)    DAC (or ZN426E-8), Ferranti or GEC Plessey 
 U10    LM339N          Quad comparator
 U11    74HC14          Hex Schmitt inverter
 U15    27C128          Eprom - TX (or 27C256 and/or NMOS)
 U16    TL084CN         Quad op-amp

 TP0-8  Test points (9)
 JMP1-5 3-pin header, male, SIL, 0.1" pitch
 JMP6-8 2-pin header, male, SIL, 0.1" pitch
    8 x  Shorting jacks - 0.1" for JMP1-8
 P1     3 pin connector, male, SIL, 0.1" pitch  )
 P2     6 pin connector, male, SIL, 0.1" pitch  )  e.g. Molex etc
 P3     5 pin connector, male, SIL, 0.1" pitch  )
 D1     1N4001 diode (or equivalent 1 amp)

 Q1     LM340T5, LM7805CT etc 5v TO-220 voltage regulator
  1 x Redpoint TV46 (TO-126 size) Heatsink. 27 degC/w, 22x19 mm
  1 x M3x6 screw and nut
 IC sockets: 2 x 28-pin.  (Optional 14 x 14-pin, 3 x 16-pin).  Turned-pin type
 1 x PCB.  Notes: Board is single Eurocard size, 160 x 100mm.  Double sided,
 plated through, max copper ground-plane, solder resist and legend.   Drilled
 for an optional DIN 41612 96 way connector. Four 3.3 mm mounting holes
 provided on 6.05 x 3.50 inch centres.  Two spare 16 pin IC positions.

 A suitable heatsink can also be fabricated from a piece of 1.6mm aluminium,
 19mm wide x 40 mm, folded into a "U" shape 19x20 and 10mm high sides, and a 3.
 3mm hole drilled in the centre.

James Miller G3RUH, 3 Benny's Way, Coton, Cambridge, CB3 7PS, England
Tel: +44 1954 210388 Fax: +44 1954 211256

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