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Spread Spectrum Voice Link

by James Vincent, G1PVZ

Adapted from an article which first appeared in ELECTRONICS + WIRELESS WORLD, Sept. and Oct. 1993. This article is reprinted with permission of the author in slightly modified form for the US and worldwide Spread Spectrum / Wireless community. Mr. Vincent brings a clear and concise style of writing to bear on a subject that is often confusing and filled with "magic." This is the first part of a two part series, which will show detailed plans and schematics for building a digital voice SS transceiver.

Due to complicated copyright issues, we are not able to post the figures for this Article. To obtain a copy of the complete article or just the figures -- please contact him directly by email:

Most Communications Engineers are used to minimizing transmission bandwidths. The trend has been to use narrower bandwidths, as with the transition from double sideband to single sideband modulation. It is quite obvious that narrower bandwidths permit more communication channels to be packed into a defined frequency band.

However the rationale of using the very wide bandwidths required by Spread Spectrum systems needs explanation. Claude Shannon produced a ground-breaking paper on the mathematical theory of communication in 1949. Shannon's resulting theorem can be expressed as:

C=W log 2 {51+S/N] bits per sec where C = data rate in bits per second; W = bandwidth (Hz), S = average signal power (W); N = mean white gaussian noise power (W). It can be seen from the equation that the only options available to increase a channel's capacity are to increase either the bandwidth (W) or the signal to noise ratio (S/N).

An increase in the signal to noise ratio requires an increase in transmitter power as the noise within the channel is beyond our control! Thus we can either trade power or bandwidth to achieve a specified channel data rate. Because of the logarithmic relationship, increasing the power output is often unrealistic. However if frequency allocation constraints permit, the bandwidth can be increased. An appreciable increase in data capacity or signal to noise ratio (for a fixed data rate) can then be achieved.

Spread spectrum systems utilize very wide bandwidths and low signal to noise ratios. From Shannon's theorem and a little algebra, we get:

C/W = 1.44 Loge [1+ S/N]

In a spread spectrum system the S/N is typically small, much less than 0.1, we get:

C/W = 1.44 S/N

From the derived relationship it can be clearly seen that a desired signal to noise ratio for a fixed data rate C, can be achieved by increasing the transmission bandwidth.

For example, assume a data rate of 32 kBits per sec and a signal to noise ratio of 0.001 (-30dB):

W=CN/1.44S thus W = 32 x 103 X 1000/1.44 + 22MHz.

So for a data rate of 32 kBits per sec, operation at the very low S/N ratio of -30db is achievable by spreading the signal over a bandwidth of 22MHz. By using a very much wider bandwidth than that of the original data it is possible to maintain data capacity without increasing the transmitter output power. It is an extreme example of a power/bandwidth tradeoff.

Two criteria (see Dixon) for a spread spectrum system are:

  • The transmitted bandwidth is much greater than the bandwidth or rate of the information being sent; and

  • Some function other than the information being sent determines the resulting radio frequency bandwidth.

The two major techniques used in spread spectrum systems are frequency hopping (FH) and direct sequence (DS). Of the two, frequency hopping is perhaps the easiest to visualize. In a frequency hopping system the frequency or channel in use is changed rapidly. The transmitter hops from channel to channel in a predetermined but pseudo-random sequence. The receiver has an identical list of channels to use (the hop set and an identical pseudo-random sequence generator to that of the transmitter. A synchronizing circuit in the receiver ensures that the pseudo-random code generator in the receiver synchronizes to the one in the transmitter. When the transmitter and receiver are synchronized the user is unaware that the transmitter and receiver are rapidly changing frequency.

However, Should the receiver not be synchronized to the transmitter or a conventional receiver be used, nothing will be heard unless the transmitter hops onto the receiver's tuned frequency. As a frequency - hopping transmitter typically hops over tens to thousands of frequencies per second (the hop rate), the time it stays on a particular channel (the dwell time) is very short and as a result the signal would appear as a burst of interference.

The other major spread spectrum technique is known as direct sequence or pseudo-noise. In this technique, a pseudorandom code directly phase shift keys the carrier increasing its bandwidth. In a typical direct sequence system, a doublebalanced mixer (DBM) is driven by the PN code to switch a carrier's phase between 0 degrees and 180 degrees. This is known as biphase shift keying (BPSK) or sometimes phase reversal keying (PRK). Unlike a frequency hopping transmitter, where the pseudo-random sequence commands a synthesizer to change frequency, the direct sequence signal is directly generated by the pseudo-random sequence.

The receiver despreads this wideband signal by using an identical synchronized pseudorandom code to that in the transmitter. As with the frequency hopper, the receiver must use a circuit to adjust its clock rate so that the receiver's pseudo-random code is at the same point in the code as the transmitter. A tracking circuit is necessary to maintain synchronism once it has been attained.

Sending data with Spread Spectrum

Spread spectrum signals ( whether direct sequence, frequency hopping or their hybrids) can support any conventional analog or digital modulation scheme to impress data onto the spread spectrum carrier.

Obviously, some modulation formats are less suitable than others. Amplitude modulation and its derivatives are the least desirable as their use will destroy the signal's uniform power spectral density. This constant carrier envelope is very desirable for spread spectrum systems designed for covert usage.

Frequency modulation (frequency shift keying for data) is often used in frequency hopping systems, but is infrequently used in direct sequence systems. This is because when a direct sequence signal passes through a squaring or frequency doubling circuit, a carrier at twice the signal's center frequency is produced. This twice frequency narrowband carrier will contain any modulation impressed on the direct sequence signal. Thus with analogue modulation it is possible for the signal to be demodulated without any prior knowledge of the pseudo-random spreading code.

One of the commonest modulation techniques used in conjunction with direct sequence is known as code inversion or modification. The digitized voice or digital data is exclusive OR'ed with the PN spreading code. This will invert the PN code sequence if the data is a "1" or pass the PN code unmodified if it is a "0". Provided that the data stream is synchronized with the PN code, the correlation properties of the code are unaffected.

Assuming synchronization at the receiver, the unmodified code despreads the direct sequence signal. This produces a narrowband signal which is still biphase shift modulated, but this time with the data or digitized speech. This signal can then be demodulated by a conventional biphase shift demodulator such as a squaring or Costas loop demodulator.

This code modification modulation is simple to implement in the transmitter and relatively easy to demodulate in the receiver. It also has the advantage of providing message privacy which the analog modulated direct sequence signal does not have. It should be noted that it is possible to directly demodulate uncorrelated spectral components of analog modulated direct sequence signal should the demodulating receiver be very close to the transmitter. In addition, the code modification technique preserves the constant power envelope of the direct sequence signal.

One disadvantage of code modification is that voice or other analog signals require digitization. As in any system design, the selection of the digitization technique is veryn important. The technique selected must use the lowest possible data rate as data rate is inversely proportional to the process gain of the system. The technique selected for the system described uses an enhanced form of delta modulation to digitally encode the voice into a serial data stream.

Delta modulation

Delta modulation is a variation of pulse-code modulation. It compares successive signal samples and transmits only their differences, rather than the actual amplitude as in PCM. This reduces the number of bits required to code the speech. The continuous audio signal is sampled at periodic intervals. The sampled value is then compared with a staircase approximation of the output signal. If the sampled waveform exceeds the staircase approximation, a positive pulse is generated. If the sampled waveform is less than the staircase approximation, a negative pulse is generated. This output pulse, positive or negative, forms the next step in the staircase approximation, i.e. the sum of the binary pulse train at the output of the encoder produces the delta modulated waveform.

At the receiver, the transmitted pulses are integrated and passed through a lowpass filter to remove unwanted high frequency components. The output consists of the original analogue signal together with some additional noise somewhat similar to quantization noise.

Continuously variable slope delta modulation (CVSD) takes advantage of the fact that voice signals do not change abruptly and that there is only a small change from one sample to the next. A reasonably good reproduction can be obtained by transmitting in a given interval whether the output signal should increase or decrease. A linear delta modulated system has the undesirable feature that there is one input level which maximizes the signal to noise ratio. In CVSD this is overcome by compressing the large amplitude in the signals relative to the smaller ones prior to encoding using a compressor circuit. In this way, the input level to the encoder can be maintained close to the value which gives the maximum signal to noise ratio.

The receiver decodes the delta modulated binary stream and passes the analog signal through an expander to counteract the effects of the transmitter compressor. Companding is optimized for the human voice. CVSD is considerably more effective than standard delta modulation and also exhibits less serious sound degradation in the presence of digital noise interference than PCM.

Circuit description

The system is described in functional blocks. First, the transmitter direct sequence modulator. The exciter's clock frequencies are provided by a master 4 MHz crystal oscillator and a divider. Power-up reset (with manual override) figured around a Schmidttrigger.

A shift register and exclusive OR gates are configured as a 4 MHz 127 chip (code bit) long maximal pseudo-random code generator.

Microphone audio is amplified by the vogad (voice operated gain adjusting device) to the optimum level for the input of the delta modulator. The delta modulator converts the audio into a 32 kBits per sec serial data stream. This serial binary data stream must be coded into a format which is polarity insensitive because the receiver demodulator cannot recover the despread data's absolute phase. Only data transitions are recovered at the receiver, hence there is no way of determining whether the output data stream is inverted or not.

The digitized audio is converted from a non return to zero (NRZ) format into a polarity insensitive diphase (biphase-mark) data stream. This subcircuit produces a diphase signal, where a logic 1 has start, mid-bit and end transitions and a logic 0 has only start and finish transitions.

In addition to providing phase insensitive data transmission, the format also makes clock recovery at the receiver relatively easy, as unlike NRZ even a continuous stream of diphase encoded 0's results in many start and finish data cell transitions. The diphase encoded delta modulated digital voice signal is EXOR'ed with the pseudo-random code producing a code modified PN spreading code.

The data modified PN code from the output of the exclusive-OR gate provides a balanced drive ( +/- 24 mA as an AC logic family device has equal sink and source currents) via a coupling capacitor and 50 ohm matching pad, to a double balanced mixer (DBM) configured as a biphase shift keyer.

The PN code output alternately sinks and sources current, causing the diodes in the DBM to alternately switch on and off producing 180 degree phase reversals in the 435 MHz carrier signal. The output spectrum consists of a series of symmetrical sidebands which have a Sinc2x distribution due to the many frequency components of the pseudorandom code. As the spreading code is pseudorandom in character, the occurrence of a particular frequency is pseudo-random in time and in the direct sequence output appears as a noise on a spectrum analyzer. The Spread spectrum signal has a main lobe bandwidth of 8 MHz (twice the PN code clock rate for BPSK). This is amplified by a MAR8 MMIC (monolithic microwave integrated circuit) and further amplified to around 100 mW by a Motorola CA4812 class A amplifier module. Helical band pass filtering is used to ensure that the output signal is within the permitted bandwidth before free-space transmission.

Spread Spectrum Terminology

Process gain (Gp) is a fundamental concept in spread spectrum systems. The process gain indicates the gain or signal to noise improvement exhibited by a spread spectrum system by nature of the spreading and despreading process. Process gain can be estimated by the following empirical relationship:

Process Gain = Gp / Rinf0 = BWRF
Process Gain = 10log10 [BWRF /Rinf0 ]dB

where BWRF is the 3dB bandwidth of the transmitted spread spectrum signal in Hz. Rinfo is the information data rate in bits per second.

For a direct sequence signal, BWRF is assumed to be equal to the 3dB bandwidth of the spectrum (which 0.8 time the pseudo-random code clock rate for a biphase shift keyed direct sequence system). For a frequency hopping system BW is equal to m times the bandwidth, where m is the number of frequency channels available.

Jamming Margin. Although the process gain is directly related to the interference rejection properties, a more indicative measure of how a spread spectrum system will perform in the face of interference is the jamming margin (Mj). The process gain of a system will always be greater than its jamming margin.

Mj = Gp - [L system - (S /N) out] dB
where Lsystem is the system implementation loss (dB); Gp is process gain (dB): (S/N) out is the signal to noise ratio at the information output (dB).

A spread spectrum system with a 30 dB process gain, a minimum required output signal to noise of 10 dB and system implementation loss of 3 dB would have a jamming margin of 30 - (10+3) dB which is 17 dB. The spread spectrum system in this example could not be expected to work in an environment with interference more than 17 dB above the desired signal.

Spectral Power Density By nature of the spread spreading process, the output power of the spectrum transmitter is spread over typically many Megahertz of bandwidth. The spectral density is the number of Watts of radio frequency power present per Hertz of bandwidth. Thus for a direct sequence transmitter of 1 W output and a spread bandwidth of 8 MHz the power spectral density is:

1 / 8,000,000 W / Hz = 125 nW / Hz

For a conventional AM transmitter, the power spectral density is around 1 /6000 W / Hz = 166 uW, some 31 dB greater.

The advantage to the military user is that the signal strength apparent to a conventional narrowband receiver is very, very low and would probably not be recognized as a communications signal, hence the expression "Low Probability of Intercept" and" Low Probability of Recognition."

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